I. Field of the Invention
The present invention relates to communications. More particularly, the present invention relates to a novel and improved programmable dynamic range receiver.
II. Description of the Related Art
The design of a high performance receiver is made challenging by various design constraints. First, high performance is required for many applications. High performance can be described by the linearity of the active devices (e.g. amplifiers, mixers, etc.) and the noise figure of the receiver. Second, for some applications such as in a cellular communication system, power consumption is an important consideration because of the portable nature of the receiver. Generally, high performance and high efficiency are conflicting design considerations.
An active device has the following transfer function: EQU y(x)=a.sub.1 .multidot.x+a.sub.2 .multidot.x.sup.2 +a.sub.3 .multidot.x.sup.3 +higher order terms, (1)
where x is the input signal, y(x) is the output signal, and a.sub.1, a.sub.2, and a.sub.3 are coefficients which define the linearity of the active device. For simplicity, higher order terms (e.g. terms above third order) are ignored. For an ideal active device, the coefficients a.sub.2 and a.sub.3 are 0.0 and the output signal is simply the input signal scaled by a.sub.1. However, all active devices experience some amount of non-linearity which is quantified by the coefficients a.sub.2 and a.sub.3. Coefficient a.sub.2 defines the amount of second order non-linearity and coefficient a.sub.3 defines the amount of third order non-linearity.
Most communication systems are narrow band systems which operate on an input RF signal having a predetermined bandwidth and center frequency. The input RF signal typically comprises other spurious signals located throughout the frequency spectrum. Non-linearity within the active devices causes intermodulation of spurious signals, resulting in products which may fall into the signal band.
The effect of second order non-linearity (e.g. those caused by the x.sup.2 term) can usually be reduced or eliminated by careful design methodology. Second order non-linearity produces products at the sum and difference frequencies. Typically, the spurious signals which can produce in-band second-order products are located far away from the signal band and can be easily filtered. However, third order non-linearity are more problematic. For third order non-linearity, spurious signals x=g.sub.1 .multidot.cos (w.sub.1 t)+g.sub.2 .multidot.cos (w.sub.2 t) produce products at the frequencies (2w.sub.1 -w.sub.2) and (2w.sub.2 -w.sub.1). Thus, near band spurious signals (which are difficult to filter) can produce third order intermodulation products falling in-band, causing degradation in the received signal. To compound the problem, the amplitude of the third-order products are scaled by g.sub.1 .multidot.g.sub.2.sup.2 and g.sub.1.sup.2 .multidot.g.sub.2. Thus, every doubling of the amplitude of the spurious signals produces an eight-fold increase in the amplitude of the third order products. Viewed another way, every 1 dB increase in the input RF signal results in 1 dB increase in the output RF signal but 3 dB increase in the third order products.
The linearity of a receiver (or the active device) can be characterized by the input-referred third-order intercept point (IIP3). Typically, the output RF signal and the third-order intermodulation products are plotted versus the input RF signal. As the input RF signal is increased, the IIP3 is a theoretical point where the desired output RF signal and the third-order products become equal in amplitude. The IIP3 is an extrapolated value since the active device goes into compression before the IIP3 point is reached.
For a receiver comprising multiple active devices connected in cascade, the IIP3 of the receiver from the first stage of active device to the n.sup.th stage can be calculated as follows: ##EQU1## where IIP3.sub.n is the input-referred third-order intercept point from the first stage of active device to the n.sup.th stage, IIP3.sub.n-1 is the input-referred third-order intercept point from the first stage to the (n-1).sup.th stage, Av.sub.n is the gain of the n.sup.th stage, IIP3.sub.dn is the input-referred third-order intercept point of the n.sup.th stage, and all terms are given in decibel (dB). The calculation in equation (2) can be carried out in sequential order for subsequent stages within the receiver.
From equation (2), it can be observed that one way to improve the cascaded IIP3 of the receiver is to lower the gain before the first non-linear active device. However, each active device also generates thermal noise which degrades the signal quality. Since the noise level is maintained at a constant level, the degradation increases as the gain is lowered and the signal amplitude is decreased. The amount of degradation can be measured by the noise figure (NF) of the active device which is given as follows: EQU NF.sub.d =SNR.sub.in -SNR.sub.out, (3)
where NF.sub.d is the noise figure of the active device, SNR.sub.in is the signal-to-noise ratio of the input RF signal into the active device, SNR.sub.out is signal-to-noise ratio of the output RF signal from the active device, and NF.sub.d, SNR.sub.in and SNR.sub.out are all given in decibel (dB). For a receiver comprising multiple active devices connected in cascade, the noise figure of the receiver from the first stage of active device to the n.sup.th stage can be calculated as follows: ##EQU2## where NF.sub.n is the noise figure from the first stage to the n.sup.th stage, NF.sub.n-1 is the noise figure of the first stage to the (n-1).sup.th stage, NF.sub.dn is the noise figure of the n.sup.th stage, and G.sub.n-1 is the accumulated gain of the first stage through the (n-1).sup.th stage in dB. As shown in equation (4), the gain of the active device can affect the noise figure of the subsequent stages. Similar to the IIP3 calculation in equation (2), the noise figure calculation in equation (4) can be carried out in sequential order for subsequent stages of the receiver.
Receivers are employed for many communication applications, such as cellular communication systems and high definition television (HDTV). Exemplary cellular communication systems include Code Division Multiple Access (CDMA) communication systems, Time Division Multiple Access (TDMA) communication systems, and analog FM communication systems. The use of CDMA techniques in a multiple access communication system is disclosed in U.S. Pat. No. 4,901,307, entitled "SPREAD SPECTRUM MULTIPLE ACCESS COMMUNICATION SYSTEM USING SATELLITE OR TERRESTRIAL REPEATERS", and U.S. Pat. No. 5,103,459, entitled "SYSTEM AND METHOD FOR GENERATING WAVEFORMS IN A CDMA CELLULAR TELEPHONE SYSTEM", both assigned to the assignee of the present invention and incorporated by reference herein. An exemplary HDTV system is disclosed in U.S. Pat. Nos. 5,452,104, 5,107,345, and 5,021,891, all three entitled "ADAPTIVE BLOCK SIZE IMAGE COMPRESSION METHOD AND SYSTEM", and U.S. Pat. No. 5,576,767, entitled "INTERFRAME VIDEO ENCODING AND DECODING SYSTEM", all four patents are assigned to the assignee of the present invention and incorporated by reference herein.
In cellular applications, it is common to have more than one communication system operating within the same geographic coverage area. Furthermore, these systems can operate at or near the same frequency band. When this occurs, the transmission from one system can cause degradation in the received signal of another system. CDMA is a spread spectrum communication system which spreads the transmit power to each user over the entire 1.2288 MHz signal bandwidth. The spectral response of an FM-based transmission can be more concentrated at the center frequency. Therefore, FM-based transmission can cause jammers to appear within the allocated CDMA band and very close to the received CDMA signal. Furthermore, the amplitude of the jammers can be many time greater than that of the CDMA signal. These jammers can cause third-order intermodulation products which can degrade the performance of the CDMA system.
Typically, to minimize degradation due to intermodulation products caused by jammers, the receiver is designed to have high IIP3. However, design of a high IIP3 receiver requires the active devices within the receiver to be biased with high DC current, thereby consuming large amounts of power. This design approach is especially undesirable for cellular application wherein the receiver is a portable unit and power is limited.
Several techniques have been deployed in the prior art to address the need for high IIP3. One such technique, which also attempts to minimize power consumption, is to implement the gain stage with a plurality of amplifiers connected in parallel and to selectively enable the amplifiers as higher IIP3 is needed. This technique is disclosed in detail in U.S. patent application Ser. No. 08/843,904, entitled "DUAL MODE AMPLIFIER WITH HIGH EFFICIENCY AND HIGH LINEARITY", filed Apr. 17, 1997, assigned to the assignee of the present invention and incorporated by reference herein. Another technique is to measure the received RF signal power and adjust the gain of the amplifiers based on the amplitude of the RF signal power. This technique is disclosed in detail in U.S. patent application Ser. No. 08/723,491, entitled "METHOD AND APPARATUS FOR INCREASING RECEIVER POWER IMMUNITY TO INTERFERENCE", filed Sep. 30, 1996, assigned to the assignee of the present invention and incorporated by reference herein. These techniques improve the IIP3 performance but have not effectively reduced power consumption nor minimized circuit complexity.
An exemplary block diagram of a receiver architecture of the prior art is shown in FIG. 1. Within receiver 1100, the transmitted RF signal is received by antenna 1112, routed through duplexer 1114, and provided to low noise amplifier (LNA) 1116. LNA 1116 amplifies the RF signal and provides the signal to bandpass filter 1118. Bandpass filter 1118 filters the signal to remove some of the spurious signals which can cause intermodulation products in the subsequent stages. The filtered signal is provided to mixer 1120 which downconverts the signal to an intermediate frequency (IF) with the sinusoidal from local oscillator 1122. The IF signal is provided to bandpass filter 1124 which filters spurious signals and downconversion products prior to the subsequent downconversion stage. The filtered IF signal is provided to automatic-gain-control (AGC) amplifier 1126 which amplifies the signal with a variable gain to provide an IF signal at the required amplitude. The gain is controlled by a control signal from AGC control circuit 1128. The IF signal is provided to demodulator 1130 which demodulates the signal in accordance with the modulation format used at the transmitter. For digital transmission such as binary phase shift keying (BPSK), quaternary phase shift keying (QPSK), offset quaternary phase shift keying (OQPSK), and quadrature amplitude modulation (QAM), a digital demodulator is used to provide the digitized baseband data. For FM transmission, an FM demodulator is used to provide the analog signal.
Receiver 1100 comprises the basic functionalities required by most receivers. However, the location of amplifiers 1116 and 1126, bandpass filters 1118 and 1124, and mixer 1120 can be rearranged to optimize the receiver performance for a particular application. In this receiver architecture, high IIP3 is provided for by biasing the active devices at high DC bias current and/or by controlling the gain of amplifier 1126.
This receiver architecture has several drawbacks. First, the active devices are typically biased to a high DC current to provide the highest required IIP3. This has the effect of operating receiver 1100 at the high IIP3 operating point at all times, even though high IIP3 is not required most of the time. Second, the high IIP3 can be improved by adjusting the gain of AGC amplifier 1126, as disclosed in the aforementioned U.S. Pat. No. 5,099,204. However, lowering the gain of amplifier 1126 can degrade the noise figure of receiver 1100.